1. Field of the Invention
The present invention relates to a synchronous rectification circuit of a DC-DC converter to commutate the voltage induces in a secondary winding of a transformer by implementing the on-off control of a commutation side field-effect transistor and a flywheel side field-effect transistor synchronously with the on-off operation of a switching transistor connected to a primary winding of the transformer.
2. Description of the Related Art
A diode has been generally employed in a commutating circuit of conventional DC-DC converters. However, the power loss due to the forward voltage drop of the diode is not negligible but large. Thus, in recent years, a Schottky diode small in power loss due to the forward voltage drop has been extensively employed.
An FET (Field-Effect Transistor) of at most several mxcexa9 in ON-resistance was developed. Thus, the power loss can be more reduced by using such an FET in a commutating circuit than by using the Schottky diode.
Thus, a configuration was proposed, in which the FET is connected to a primary winding of a transformer as a switching transistor, and is used in the commutating circuit of the DC-DC converter. In this case, the FET as the switching transistor implements the on-off control of the FET of the commutating circuit in a synchronous manner with the on-off timing. The commutating circuit can thus commutate the induced voltage in the secondary winding of the transformer with less power loss.
FIG. 1 is a circuit diagram of a conventional DC-DC converter, indicating a synchronous rectification circuit applied to a forward converter. In FIG. 1, T denotes a transformer, n1 denotes a primary winding, n2 denotes a secondary winding, Q1 denotes a switching transistor (FET, Q2 denotes a commutating side FET, Q3 denotes a flywheel side FET, and L and C1 denote a reactor and a capacitor to constitute a smoothing circuit, respectively. A control circuit 1 detects the output voltage Vout, and controls the ON-period of the switching transistor Q1 by controlling the pulse width or the like. This means that the control circuit 1 controls the ON-period and the OFF-period of the switching transistor Q1 so that the smoothed output voltage Vout is the preset voltage. More specifically, the switching transistor Q1 turns on/off the current flowing in the primary winding n1 of the transformer T based on the input voltage Vin. The control circuit 1 shortens the ON-period when the output voltage Vout is higher than the preset voltage, and extends the ON-period when the output voltage Vout is lower than the preset voltage. The control circuit 1 controls the output voltage Vout to be constant by the above operation. The detailed operation is described below.
When the switching transistor Q1 is turned on, the commutating side FET Q2 is turned on by the induced voltage in the secondary winding n2 of the transformer T. As a result, the current flows in the capacitor C1 via the reactor L. In this condition, the flywheel side FET Q3 is turned off.
When the switching transistor Q1 is turned off, the polarity of the induced voltage in the secondary winding n2 of the transformer T is inverted, and the commutating side FET Q2 is turned off. As a result, the flywheel side FET Q3 is turned on, and the current attributable to the accumulated energy in the reactor L flows in the capacitor C1.
However, the transformer T is reset (to make the accumulated energy in the transformer T zero) through the resonance effect of the parasitic capacitance or the like or the switching transistor Q1 with the inductance of the transformer T. After the transformer T is reset, the induced voltage in the secondary winging n2 becomes zero. This means that the voltage applied to the gate of the flywheel side FET Q3 becomes zero during the OFF-period of the switching transistor Q1, and the flywheel side FET Q3 is turned off. Thus, the flywheel current flows via a parasitic diode (a body diode) of the flywheel side FET Q3, raising a problem of not taking advantage of the low ON-resistance of the FET.
Thus, the DC-DC converter shown in FIG. 2 was proposed. In the figure, the same symbols as those in FIG. 1 show the same parts. In FIG. 2, n3 denotes a tertiary winding of a transformer T, CT denotes a current transformer, Q4 denotes a transistor, D7 and D8 denote diodes (body diodes), Cgs denotes the gate-source parasitic capacitance (input capacitance), and Cgd denotes the gate-drain parasitic capacitance (input capacitance), respectively. The control circuit 1 used to implement the on-off control of the switching transistor Q1 by detecting the output voltage Vout is omitted in the figure.
The tertiary winding n3 is provided on the transformer T, and the induced voltage in this tertiary winding n3 is applied to the gate of the commutating side FET Q2. When the switching transistor Q1 turned on/off thereby, the commutating side FET Q2 is turned on/off in a synchronous manner with the on-off operation of the switching transistor Q1.
The primary winding of the current transformer CT is connected in series to the flywheel side FET Q3, and the induced voltage in the secondary winding is applied to the gate of the flywheel side FET Q3 via the diodes D7 and D8. The resistor R3 corresponds to the terminating resistor of the current transformer CT. The induced voltage in the secondary winding of the current transformer CT is applied to the Zener diode ZD7. The Zener diode ZD7 suppresses the voltage across the resistor R3 to the Zener voltage, and applies it to the base of the transistor Q4. Further, the Zener diode ZD7 suppresses the voltage across the resistor R3 to the Zener voltage, and applies it to the gate of the flywheel side FET Q3 via the diode D8.
When the switching transistor Q1 on the primary side of the transformer T is turned on, the commutating side FET Q2 is turned on by the induced voltage in the tertiary winding n3 of the transformer T. As a result, the current by the induced voltage in the secondary winding n2 flows via the turned-on commutating side FET Q2. In this condition, the flywheel side FET Q3 is turned off.
When the switching transistor Q1 is turned off, the polarity of the induced voltage in the tertiary winding n3 of the transformer T is inverted, and the commutating side FET Q2 is turned off. The flywheel current flows via the body diode Dq3 of the flywheel side FET Q3. This current flows in the primary winding of the current transformer CT. As a result, the induced voltage in the secondary winding of the current transformer CT is applied to the gate of the flywheel side FET Q3 via the diodes D7 and D8 to charge the input capacitances Cgs and Cgd. When this charging voltage exceeds the threshold, the flywheel side FET Q3 is turned on. Even when the induced voltage in the secondary winding n2 becomes completely zero by resetting the transformer T, the flywheel side FET Q3 continues the ON condition thereof by the induced voltage in the secondary winding of the current transformer CT.
FIG. 3 shows the flywheel side FET Q3 illustrated in FIG. 2 as a three-terminal element. FIG. 4, similar to FIG. 3, shows the flywheel side FET Q3 illustrated in FIG. 2 as a two-terminal element. More specifically, the induced voltage in the secondary winding of the current transformer CT is applied to the gate of the flywheel side FET Q3. This means that the flywheel side FET Q3 is turned off by the induced voltage in the secondary winding of the current transformer CT if the forward, current flows in the body diode thereof. Thus, in this case, the flywheel side FET Q3 can be used for the two-terminal element diode of the low voltage drop characteristic as illustrated in FIG. 4.
FIG. 5 is a schematic representation of the current waveform and the voltage waveform on the flywheel side FET Q3 shown in FIG. 2.
In FIG. 5, Id denotes the current flowing in the primary winding of the current transformer CT, Vgs denotes the gate-source voltage, and Vds denotes the drain-source voltage. TS denotes the period of the on-off control, Ton and Toff denote the ON-period and OFF period of the switching transistor Q3, Vf denotes the forward voltage of the body diode Dq3, and Td denotes the current flow period of the body diode Dq3.
When the switching transistor Q1 is turned on, the flywheel side FET Q3 is turned off as described above, and the drain-source voltage Vds is increased. When the switching transistor Q1 is turned off, the current flows via the body diode Dq3 of the flywheel side FET Q3 after a specified time is elapsed. Thus, the drain-source voltage Vds becomes the forward voltage Vf of the body diode Dq3. As illustrated in FIG. 5, the flywheel side FET Q3 is turned on by the induced voltage in the secondary winding of the current transformer CT after the period Td is elapsed, and the drain-source voltage Vds thereof becomes substantially zero.
FIG. 6 is a measured waveform chart to show the operation of the prior art technology shown in FIG. 2. In FIG. 6, Q1Vds and Q1Id denote the drain-source voltage and the drain current of the switching transistor Q1, and Q3Vgs and Q3Vds denote the gate-source voltage and the drain-source voltage of the flywheel side FET Q3. Td denotes the current flow period in the body diode Dq3 of the flywheel side FET Q3. As illustrated in FIG. 6, the current flow period Td in the body diode Dq3 of the flywheel side FET Q3 is relatively long as described in relation to FIG. 2.
The prior art technology shown in FIG. 2 involves the following problems.
In the process to turn on the flywheel side FET Q3, a specified time is required for the rise of the gate-source voltage Vgs. Thus, the loss in the current flow period Td in the body diode Dq3 becomes a problem.
To shorten the above period Td, the following countermeasures may generally be taken. The current transformer CT which can rapidly charge the input capacity. Cgs of the flywheel side FET Q3 is used so that the gate-source voltage Vgs may be charged in a short time.
However, if such a current transformer CT is used, the current flowing in the secondary winding of the current transformer CT is increased, and the driving loss of the flywheel side FET Q3 is increased, raising a problem that the driving loss of the synchronous rectification circuit of the DC-DC converter can not be reduced.
Accordingly, a first object of the present invention is to reduce the driving loss (the power loss) of a synchronous rectification circuit of a DC-DC converter with a relatively simple configuration.
A second object of the present invention is to increase the speed of the turning on of a flywheel side FET included in the synchronous rectification circuit, and to reduce the loss by a body diode.
A third object of the present invention is to reduce the driving loss after the flywheel side FET included in the synchronous rectification circuit is turned on.
A fourth object of the present invention is to prevent the DC-DC converter in the light-load mode from being inoperable due to the driving loss of the flywheel side FET included in the synchronous rectification circuit.
A fifth object is to prevent the ratio of the ON-period to the OFF-period of a switching transistor provided on the primary side of a transformer from being limited. In other words, a current transformer included in the synchronous rectification circuit must usually prevent the saturation of the core thereof. For this purpose, the ratio of the ON-period to the OFF-period of the switching transistor is limited so that the product of the current and the time corresponding to the ON-period of the switching transistor is substantially equal to the product of the current and the time corresponding to the OFF-period of the switching transistor.
The first to third objects are achieved by changing the current transformation ratio of the current transformer included in the synchronous rectification circuit as described below, wherein the current transformation ratio means the ratio of the current I1 flowing in the primary winding to the current I2 flowing in the secondary winding of the current transformer (I1/I2).
According to one of the aspects of the present invention, the present invention includes a current controlling part for setting the current transformation ratio of the above current transformer to be a small value during the initial period when the flywheel side FET is turned on, and sets the current transformation ratio of the current transformer to be a large value after the above-described initial period has elapsed.
The input capacitance of the flywheel side FET is rapidly charged, and the turn-on speed of the flywheel side FET is increased. As a result, the driving loss attributable to the body diode of the flywheel side FET is reduced. The driving loss is also reduced after the flywheel side FET is turned on.
The fourth object of the present invention can be achieved in the following way.
According to another aspect of the present invention, it is possible to omit a terminating resistor on the secondary side of the saturable current transformer because the saturable current transformer is used for a current transformer of the synchronous rectification circuit. Since power consumption by the terminating resistor is eliminated, cases can be prevented where the flywheel side FET cannot be driven in a light-load mode.
The fifth object of the present invention can be achieved in the following way.
According to another aspect of the present invention, the saturable current transformer induces the voltage in the secondary winging and the current flows, only in the initial period when the switching transistor provided on the primary side of the transformer turns on/off. Thus, the ratio of the ON-period of the switching transistor to the OFF-period can be arbitrarily controlled. Thus, a problem of the core saturating because of the long ON-period of the FET by using a regular current transformer can be solved by using the saturable current transformer.